Direct conversion tuner

ABSTRACT

A direct conversion tuner for tuning either analog or digital television signals includes a first and second channels, each having first and second mixers and an intervening filter stage, coupled between an RF input and an output combining unit. The first mixers receive respective first local oscillator signals which have the same frequency but a quadrature phase relationship. The frequency of the first local oscillator signals is controlled according to the selected channel so that it is located within the spectrum of the respective RF signal. The second mixers receive respective second local oscillator signals which have the same frequency but a quadrature phase relationship. The frequency of the second local oscillator signal is located above the passband of the filter stages. A digital gain and phase equalization network is included in one of the channels for adjusting the relative gain and phase shift of the two channels and is automatically controlled by a microcomputer in response to signals sampled at respective points within the first and second channels to reduce the relative gain and phase shift. As a result, near perfect cancellation of unwanted components occurs in the output combining unit.

This is a continuation of application Ser. No 08/553,264, filed Nov. 22,1995 now abandoned, which is a 371 of PCT/IB94/00138, filed Jun. 6,1994.

FIELD OF THE INVENTION

The invention concerns a so called "direct conversion tuner" which isespecially useful in a television receiver.

BACKGROUND OF THE INVENTION

An early type of tuner known as a "tuned radio frequency tuner" (TRF)included several radio frequency (RF) amplifiers which were all tuned tothe frequency of the RF signal of desired transmission channel followeddirectly by a detection section, without an intervening mixer employedin later tuners. Such a tuner could provide relatively distortion freeperformance due to the absence of a mixer. However, TRF tuners tended tobe large in size and subject to stability and gain control problems dueto the number RF amplifiers which were needed. Moreover, TRF tuners didnot provide a consistent or adequate degree of signal selectivity.

The type of tuner which is primarily used today is known as a"heterodyne" or "superheterodyne" tuner. In its simplest form, known asa "single-conversion" tuner, it comprises a tunable RF amplifierfollowed by a frequency conversion stage, including a mixer and a localoscillator. The frequency conversion stage produces an intermediatefrequency (IF) signal which corresponds to the received RF signal buthas a lower frequency. The IF signal is filtered by an IF filter sectionand the resultant signal is coupled to a detection section. Thecombination of conversion stage and the following IF filter sectionprovides a significantly better selectivity characteristic than a TRFtuner. The frequency of the local oscillator signal is offset (usuallyhigher) from the frequency of the desired RF signal by the desiredfrequency of the IF signal. In a television receiver, the localoscillator signal is controlled so that it places the frequency of theIF picture carrier corresponding to the RF picture carrier at a nominalfrequency, for example, at 45.75 MHz in the United States and 38.9 MHzin Europe.

A single conversion tuner can be made quite small and relativelyinexpensive. However, it produces unwanted intermodulation andcross-modulation products due to third and higher order components ofthe signal transfer characteristics of the mixer included in thefrequency conversion stage. Various unwanted conversion products, knownin the tuner fields as "image", "one-half IF" and "IF beats", continueto be a problem. The IF filter is designed to minimize unwantedconversion products and also to provide rejection of responses due toadjacent channels (selectivity). Thus, the selection of the IF frequencyis a compromise. As a result the rejection of unwanted conversionproducts and selectivity of the tuner may not be adequate.

The deficiencies of a single conversion tuner have become especiallytroublesome due to the increasing number of "contiguous" channels nowavailable in large cable television systems. With the advent of digitaltelevision transmission systems, such as for high definition television(HDTV), the problem becomes still more difficult because these systemsmake full use of the available channel spectrum and only a small guardband of a few hundred kilohertz (KHz) exists between channels. Inaddition, the overall frequency response of a tuner for tuning digitaltelevision signals must be flat to the edges of the channel, butnevertheless, have a very steep "roll-off" (attenuation) at the edgesfor adequate adjacent channel rejection. This makes the design of anappropriate IF filter more complicated since no Nyquist slopes and soundtraps, which tend to ease IF filter design, can be used in digitalsystems. In addition, it is contemplated that both analog and digitaltelevision signals will be transmitted during a transition period. Inthat case, even more adjacent channel selectivity will be required forgood reception of the digital signals because digital television signalswill be transmitted with much less power than analog television signals.

The "double-conversion" variation of the superheterodyne tuner wasdeveloped to overcome the shortcomings of the single-conversion tuner.In this type of tuner, a first conversion stage is followed by a firstIF filter section, a second conversion stage, and a second IF filterstage. The first IF section has a very high frequency range, typicallyin the order of 620 MHz. The second IF section has a much lowerfrequency range, typically the same as that of the only IF filtersection of a single conversion tuner. The second IF section is followedby a detection section.

The very high frequency of the first IF filter section places the RFsignals corresponding to unwanted conversion responses such as the"image" response at frequencies readily rejected by tunable RF stageswhich precede the first conversion stage. The low frequency second IFprovides the required adjacent channel selectivity needed for moderntelevision reception. Unfortunately, an double-conversion tuner systemrequires additional RF and IF circuitry compared to a single conversiontuner, and much of the additional circuitry must function at relativelyhigh frequencies requiring extensive shielding. As a result a doubleconversion tuner is relatively large in size and expensive.

Another type of tuner known as a "direct conversion" tuner has improvedunwanted conversion product rejection and selectivity properties withrespect the TRF and heterodyne types of tuners. A direct conversiontuner operates in accordance with a third tuning method in which thefrequency of a local oscillator signal of a first frequency conversionstage is set in the middle of the frequency band of the desired channel.The product of the first conversion stage is at relatively lowfrequency. There are no image responses because the frequency of thefirst conversion stage is located with the spectrum of the desired RFsignal. In addition, the very low frequency range of the signal producedat the output of the first conversion stage makes it possible to readilyprovide a filter which can reject adjacent channel signals.

Unfortunately, because the first local oscillator signal is centered inthe frequency band of the desired channel, both the upper and lower sideband of the desired channel will be converted to the frequency range ofthe first IF signal so that the lower side band (LSB) is in effectfolded over onto the upper side band (USB) in the spectrum of the firstIF signal. Since the LSB and USB occupy the same frequency range, theLSB and the USB must again be separated before detection. To accomplishthis, a direct conversion tuner is arranged as is shown in FIG. 1.

Basically, the direct conversion tuner contains two channels, each withtwo conversion stages. The received RF signal is coupled to each of twomixers M1A and M1B via a tuned RF amplifier which provides gain and someselectivity. Desirably, the gain of the RF amplifier is automaticallycontrolled in response to an automatic gain control (AGC) signal (notshown). The local oscillator signal generated by a first localoscillator LO1 is tuned to the center frequency too of the frequencyband of the desired channel between the lower sided band (LSB) and theupper side band (USB), as is shown in FIG. 2a. The first localoscillator signal is split by a phase shifting circuit PS1 intoquadrature components that are used to drive mixers M1A and M1B. Therespective IF output signals of mixers M1A and M1B are filtered by twolow pass filters LPF A and LPF B. Low pass filters LPF A and LPF Bprovide the necessary selectivity to reject the responses from theadjacent channels and higher order products of mixers M1A and M1B.

Each of the output signals of mixers M1A and M1B includes both a lowerside band portion and an upper side band portion corresponding to theLSB and USB portions of the received RF signal. However, as earlierindicated, the LSB portion is folded over so that it is superimposed onthe USB portion and occupies the same frequency range, as is shown inFIG. 2b. The output signal of low pass filters LPF A and LPF A arecoupled to respective ones of a second pair of mixers M2A and M2B.Mixers M2A and M2B are driven by respective ones of a second pair ofquadrature local oscillator signals generated by a second localoscillator LO2 and a second phase shifting circuit PS2. Each of thesecond local oscillator signals has a frequency ω_(N) located above thecutoff frequency of the low pass filters LPF A and LPF B filters tofulfill the Nyquist criteria. The output signals of mixers M2A and M2Bare added in a summer unit SU to produce an output signal which has aspectrum which includes separated lower and upper side band portions, asshown if FIG. 2c. This output signal is coupled to a demodulator (notshown) which demodulates it, and the demodulated resultant is coupled tofurther signal processing sections.

The operation of the direct conversion tuner shown in FIG. 1 can bemathematically understood by considering a very simple case in which thereceived RF signal is assumed to include a sinusoidal upper side bandcomponent of sin (ω₀ +ω₁) and a sinusoidal lower side band component ofsin (ω₀ -ω₂), as is indicated in FIG. 3a. It is also assumed that thegains and phase shifts of the two channels are identical. The phases ofthe signal components produced at various point of the direct conversiontuner are indicated by the vector arrows in FIG. 1. Further, thecoefficients of the various mathematically factors corresponding tosignal components have been normalized in the following description.

The quadrature first local oscillator signals applied to first mixersM1A and M1B are expressed as sin ω₀ and cos ω₀, respectively; and thequadrature second local oscillator signals applied to second mixers M2Aand M2B are expressed as sin ω_(N) and cos ω_(N), respectively. Thefollowing signal is produced at the output of low pass filter LPF A:

    cos ω.sub.1 +cos ω.sub.2

The following signal is produced at the output of low pass filter LPF B:

    sin ω.sub.1 -sin ω.sub.2

The spectra at the outputs of low pass filters LPF A and LPF B are shownin FIG. 3b.

The result of the second mixing operation by mixer M2A produces thefollowing output signal:

    sin (ω.sub.N +ω.sub.1)+sin (ω.sub.N -ω.sub.1)+sin (ω.sub.N +ω.sub.2)+sin (ω.sub.N -ω.sub.2)

The result of the second mixing operation by mixer M2B produces thefollowing output signal:

    sin (ω.sub.N +ω.sub.1)-sin (ω.sub.N -ω.sub.1)-sin (ω.sub.N +ω.sub.2)+sin (ω.sub.N -ω.sub.2)

The addition of the two output signals of mixers M2A and M2B by summerSU results in the following signal:

    sin (ω.sub.N +ω.sub.1)+sin (ω.sub.N -ω.sub.2)

The spectrum at the output of summer SU is shown in FIG. 3c.

The operation of the direct conversion tuner depends on the cancellationof unwanted components developed in the two channels (compare the outputsignals of mixers M2A and M2B indicated above including the terms sin(ω_(N) -ω₁) and sin (ω_(N) +ω₂)). As was stated above, the descriptionof the operation of the direct conversion tuner so far provided assumesthat the gains and phase shifts of corresponding elements of the twochannels are identical, resulting in perfect cancellation of theunwanted components after the addition of the output signals of the twochannels by summer SU. However, in practice, gain and phasecharacteristics of the two channels are unequal and change withtemperature and time. The gain and phase characteristics affect thephase and magnitude of the vectors shown in FIG. 1. As a result, perfectcancellation of the unwanted components no longer occurs causing thegeneration of unwanted spurious components in the output signal producedby summer SU and the reduction of the quality of the demodulated signal.This is especially the case when the received RF signal is relativelycomplex, such as a television signal, and does not simply contain alower and an upper sinusoidal component as assumed in the abovedescription.

The generation of unwanted spurious components when a television signalis tuned by a direct conversion tuner of the type shown in FIG. 1 isillustrated in FIGS. 4a, 4b and 4c. FIG. 4a shows the spectrum of atelevision signal of a single channel. It includes a picture carrier(PIX), a color subcarrier (SC) and a sound carrier (SOUND). Thefrequency, ω₀, of the first local oscillator signal is locatedapproximately midway between the picture carrier and the sound carrier.FIG. 4b shows the spectrum of the signal resulting from the first mixingoperation. FIG. 4c shows the spectrum of the output signal of summer SU.For each desired component of the output signal of summer SU to theright of the frequency, w_(N), of the second local oscillator signal, anundesired "companion" to the left exists; and for each desired componentof output signal of summer SU to the left of frequency ω_(N) anundesired "companion" to the right exists. For instance, a "companion"of the picture carrier is present to the right of ω_(N) between thecolor subcarrier and the sound carrier. The presence of the unwanted"companions" causes annoying beat patterns in the demodulated videosignal and may also adversely affect the demodulated sound signal. Suchunwanted components should desirably be suppressed in the order of 45 to50 dB for optimum performance of the television receiver. This meansthat the gain and phase errors should desirably be kept less than 0.05dB. and 0.5 degrees, respectively, for optimum performance of thetelevision receiver. Such performance standards cannot be obtained andmaintained with manual adjustments.

SUMMARY OF THE INVENTION

The present invention concerns an arrangement for automatically reducingthe gain and phase difference errors of the two channels of a directconversion tuner in order to reduce the generation of unwantedcomponents in the output signal. In accordance with an aspect of theinvention, a multiple frequency reference signal is inserted as a testsignal at respective insertion points of the two channels and signalsproduced at respective measurement points of the two channels arecompared as to amplitude and phase to produce so called "ripple" gainand phase difference responses related to the gain and phase shiftdifferences between two channels, for example, due to the IF low passfilters. In accordance with another aspect of the invention, the signalsdeveloped at the measurement points in response to the RF signal for theselected channel are also compared as to amplitude and phase to produceso called "DC" gain and phase difference values related to gain andphase shift differences of one or more of the conversion stages. The"ripple" gain and phase responses and the "DC" gain and phase differencevalues are used to control a gain and phase correction network so thatthe amplitude and phase differences between the measured signals of thetwo channels are reduced. Preferably, the arrangement is implemented ina digital embodiment in which analog-to digital converters are includedin respective ones of the channels after the first conversion stages andthe gain and phase correction units comprise digital filters after thesecond conversion stages. The operation of automatic gain and phaseequalization arrangement can be automatically initiated each time thetelevision receiver is switched on or a new channel is selected.

These and other aspects of the invention will be described with respectto the accompanying Drawings.

BRIEF DESCRIPTION OF THE DRAWING

In the Drawing, FIGS. 1, 2a-2c, 3a-3c and 4a-4c concern the backgroundof the invention and have previously been described. Briefly:

FIG. 1 is a block diagram of a direct conversion tuner as is known inthe prior art;

FIGS. 2a, 2b and 2c are graphical representations of the spectra ofsignals at various points of the direct conversion tuner shown in FIG.1;

FIGS. 3a, 3b and 3c are graphical representations of the spectra ofsignals at various points of the direct conversion tuner shown in FIG. 1assuming that the input signal consists of two sinusoidal components;and

FIGS. 4a, 4b and 4c are graphical representations of the spectra ofsignals at various points of the direct conversion tuner shown in FIG. 1when the input signal comprises an analog television signal.

The remaining Figures of the Drawings concern the embodiment of theinvention. Briefly:

FIG. 5 is a block diagram of a direct conversion tuner including anautomatic gain and phase equalization arrangement constructed inaccordance with an aspect of the invention;

FIG. 6 is a vector diagram useful in understanding how gain and phaseerrors are measured by the automatic gain and phase equalizationarrangement;

FIGS. 7a and 7b, 8a and 8b, and 9a and 9b are graphical representationsof spectra of phase and gain errors useful in understanding how phaseand gain correction information used by the automatic gain and phaseequalization arrangement is obtained;

FIG. 10 is a vector diagram also useful in understanding how phase andgain correction information used by the automatic gain and phaseequalization arrangement is obtained;

FIGS. 11a and 11b are graphical representations of spectra of phase andgain correction responses needed for phase and gain equalization whichresult from the operations illustrated in FIGS. 7a and 7b, 8a and 8b, 9aand 9b and 10;

FIG. 12 is a flow chart indicating the overall operation of theautomatic gain and phase equalization arrangement.

DETAILED DESCRIPTION OF THE DRAWING

The direct conversion tuner shown in FIG. 5 is generally similar to theone shown in FIG. 1, but includes additional elements which comprise anautomatic gain and phase equalization arrangement constructed inaccordance with an aspect of the invention and certain related elements.Those elements of the direct conversion tuner shown in FIG. 5 which havethe same or similar functions as corresponding elements of the directconversion tuner shown in FIG. 1 are identified by the same or similarreference designations and will not be described in details again.

The automatic gain and phase equalization arrangement comprises a gainand phase correction network including a first gain control unit labeledGC and a second gain control unit which is actually used for phasecorrection and is therefore labeled PC. The gain and phase correctionnetwork is inserted in channel A between LPF A and output summer SU2.Gain correction unit GC and phase correction unit PC comprise respectiveprogrammable digital filters which are controlled by a microcomputer MCto adjust the gain and phase characteristics of channel A so that thegain of phase characteristics of the channels A and B are substantiallyidentical. Gain correction unit GC and phase correction unit PC may, forexample comprise, finite impulse response (FIR) filters. MicrocomputerMC samples the signals developed at points A and B just before secondmixers M2A and M2B in response to a test signal and determines therelative amplitudes and phases of the sampled signals to develop filtercoefficient control signals for gain correction unit GC and phasecorrection unit PC. For this purpose, microcomputer MC generates areference signal labeled ω_(REF) which is inserted as a test signal justafter first mixers M1A and M1B, for example, via resistors RA and RB.Microcomputer MC also samples the signals developed at points A and B inresponse to the received RF signal for the selected channel anddetermines the relative amplitudes and phases of the sampled signals theto develop the filter coefficient control signals. The generation of thefilter coefficient control signals will be described in detail below.Microcomputer may comprise the same microcomputer which is used tocontrol other functions of the television receiver.

In the embodiment of invention shown in FIG. 5, the gain and phasecorrection is accomplished by the addition of two signals which are inquadrature phase relationship as will be described below in greaterdetail with respect to the vector diagram shown in FIG. 10. For thispurpose gain correction unit GC and phase correction unit PC (which isactually another gain control unit) are included in separate paths forwhich the respective signals are in quadrature phase relationship. Morespecifically, gain correction unit GC receives a signal developed at apoint C at the output of second mixer M2A while phase correction unit PCreceives a signal developed at a point F at the output of an additionalmixer M3. The same local oscillator signal which is coupled to secondmixer M2B of channel B is coupled to additional mixer M3 so thatoscillator signals coupled to mixers M2A and M3 have a quadrature phaserelationship (sin ω_(N) and cos ω_(N), respectively). As a result, thesignals developed at points C and F have a quadrature phaserelationship. The output signal of gain control unit GC developed at apoint C' is added to the output signal of phase correction unit PCdeveloped at a point F' by an additional summer unit SU2. The signalproduced at a point G at the output of additional summer unit SU2 iscombined with to the signal produced at a point D at the output of amixer M2B by a summer unit SU1 corresponding to summer unit SU of thedirect conversion tuner shown in FIG. 1. The signal developed at a pointE at the output of summer SU1 is coupled to a demodulator section (notshown).

Since the gain and phase equalization apparatus is implemented indigital form in the direct conversion tuner shown in FIG. 5,analog-to-digital converters ADC A and ADC B have been added torespective ones of channels A and B prior to the second conversionstages. The ADC A and ADC B receive amplified versions of the outputsignals of LPF A and LPF B from respective ones of amplifiers AMP A andAMP B and provide respective digital versions of the output signals ofLPF A and LPF B. The outputs of amplifiers AMP A and AMP B arecapacitively coupled to the inputs of the ADC A and ADC B via respectiveones of capacitors CA and CB to avoid DC drift problems. The amplifiersmay be omitted when the outputs signals of LPF A and LPF B havesufficient amplitudes to permit reliable analog-to-digital conversion.In the direct conversion tuner shown in FIG. 5, second mixers M2A andM2B comprise digital multipliers instead of analog mixers and summersSU1 and SU2 are digital adders.

Before describing the gain and phase equalization operation in detail itis helpful to describe the nature of the gain and phase errors. The gainand phase errors can be divided into "ripple" and "DC" errors. The"ripple" error is caused by the differences between the gain and phaseresponses of low pass filters LPF A and LPF B. It varies as a functionof the IF frequency, but is constant for all received televisionchannels. The "DC" error is caused by quadrature errors and gaindifferences between first mixers M1A and M2B. It is constant for aselected channel but varies with the frequency of the first localoscillator. Second mixers M2A and M2B do not introduce additional errorsif they are implemented in digital form as is the case in the directconversion tuner shown in FIG. 5.

There is an additional error which can be referred to as an "asymmetry"error which occurs when first mixers M1A and M1B are implemented as adoubly-balanced mixer. The use of a doubly-balanced mixer is desirablesince it tends to reduce the coupling of the RF signal and localoscillator signal to the output of the mixer because of its balancedconfiguration. However, a doubly balanced mixer is not perfectlybalanced and this results in slightly different gain and phasecharacteristics for RF signal components which have a frequency belowthe frequency, ω₀, of the first local oscillator signal than for RFsignal components which have a frequency above the frequency of thefirst local oscillator signal. The "asymmetry" error is small and can beneglected in most cases. However, it can be corrected, if desired, aswill be described below.

It is possible to measure gain and phase errors with the use of a"sweep" generator which is connected to the antenna input, and which"sweeps" the frequency range of a selected channel. This method isrelatively complicated and expensive. Measuring the ripple and DC errorsseparately is simpler.

The ripple error can be measured by using a multiple frequency referencesignal, labeled ω_(REF) in FIG. 5. The reference signal has a lowfrequency range which only needs to be sufficiently broad to cover theIF frequency range. Experiments have shown that eight to ten discretetest frequencies are sufficient to obtain satisfactory results. Thereference signal may be generated by an oscillator (not shown) under thecontrol of microcomputer MC or directly by microcomputer MC by using alook-up table. During the ripple error test, the tuner has to bedisabled from responding to the received RF signal. This may beaccomplished by turning off the first local oscillator (LO1) ordisabling the RF stage. The ripple errors can be measured each time thetelevision is switched on or when a new channel is selected. Once theripple errors have been measured, the normal operation of the tuner isagain initiated.

The DC errors of mixers M1A and M1B are local oscillator orchannel-dependent, but are otherwise constant over the frequency rangeof each channel. Therefore, the DC errors can be measured using a singlefrequency test signal for each selected channel. However,advantageously, the picture and sound carriers of the receivedtelevision signal can be used to measure the DC errors. Both carriersare known in frequency and have high energy which makes the measurementreliable. Using both the picture and sound carriers allows for thecorrection of the asymmetry errors because, as shown in FIG. 4a, thefrequency of the picture carrier is below the frequency, ω₀, of thefirst local oscillator signal and the frequency of the sound carrier isabove the frequency of the first local oscillator signal. The carriersare measured at points A and B at the outputs of ADC A and ADC B duringthe occurrence of the broad vertical equalizing pulses when the picturecarrier has its highest energy but is not modulated with videoinformation. However, because both picture and sound carriers arepresent, the carriers need to be separated. The separation can beaccomplished after the measurement by microcomputer MC in accordancewith a software filter program. For example, low pass and high passfilter responses can be obtained utilizing a MatLab™ programcommercially available from MATHWORKS, Inc., of Massachusetts. The lowand high pass filter responses can be used to separate the picturecarrier and sound carrier responsive portions of the measurement values.Alternatively, microcomputer MC may comprise a digital signal processor(DSP) unit including digital filters and a microprocessor mayadvantageously be used. In that case, the digital filters can be used toseparate the picture and sound carriers.

The phase error for a particular test can be calculated by consideringthe vectors "a" and "b" representing the digitized signals measured atpoints A and B and the vector "c" representing their differences as thethree sides of a triangle, as is shown in FIG. 6, and by using thegeometric cosine law to find the angle y between vectors "a" and "b".The gain error is the ratio of the magnitudes of vectors "a" and "b".This is valid only for RMS (root mean square) values so that a number ofsamples is required. More specifically, for each sample, the signallevels (A and B) at points A and B are measured and squared (A² and B²)by microcomputer MC. In addition, for each sample, the differencebetween the signal levels (A-B) at points A and B is calculated andsquared ((A-B)²). The squared values for all of the samples are added.The squared magnitude of third side "c" of the triangle is related tothe sum of the squared values of the difference values (Σ(A-B)²). Thesquare root of the ratio of the sums of the squared signal values ((ΣA²/ΣB²)^(1/2)) is proportional to the relative gains of the two channels.The cosine law equation indicated in FIG. 6 is used to calculate therelative phase angle g from the appropriate sums of squared values. Arelatively large number of samples, for example, in the order of 500 ormore, because the magnitudes of the signals measured at points A and Bare continuously changing, especially during the DC and asymmetry error-measurements when the measured signals are responsive to the carriersof the received RF signal. This method provides sufficient accuracy inan eight-bit environment.

The use of the cosine law in the above described manner does not resultin reliable results for very small phase difference angles. The problemis solved in software by means of the addition of a phase shift to oneof the signals measures at points A and B. The phase shift is lattersubtracted from the calculated phase difference.

Once the gain and phase error values have been measured and stored inmicrocomputer MC, the necessary coefficients for the correction filterscan be calculated. FIGS. 7a and 7b show examples of measured ripplephase and gain errors for one half of the received spectrum. The phaseand gain errors for the other half of the spectrum are obtained byforming the mirror image of the existing phase error about the zeropoint and the mirror image of the existing gain error about the verticalaxis. The resulting full spectra of the ripple phase and gain errors areshown in FIGS. 8a and 8b.

The ripple phase and gain error responses have to be combined with theDC phase and gain error responses. More specifically, the DC phase erroris added to the ripple phase error response shown in FIG. 8a. The DCgain error is used to multiply the ripple gain error response shown inFIG. 8b.

If asymmetry errors exists, they can be compensated by using theasymmetry errors measured by using the picture and sound carriers, asearlier indicated. More specifically, the asymmetry phase error for thepicture carrier is added to the left side of the ripple phase errorresponse shown in FIG. 8a and the asymmetry phase error for the soundcarrier is added to the right side. The asymmetry gain error for thepicture carrier is used to multiply the left side of the ripple gainerror response shown in FIG. 8b and the asymmetry gain error for thesound carrier is used to multiply the right side. As a result of thecombination of the asymmetry errors with the ripple errors, a step inthe middle of the responses may occur. In many cases the asymmetryerrors are small and can be ignored, and therefore only the response tothe picture carrier is needed. In the present example, it is assumedthat the asymmetry errors are small, and therefor, have been ignored.

The final phase and gain error responses are shown in FIGS. 9a and 9b.The phase and gain error responses shown in FIGS. 9a and 9b cannot bythemselves be used for calculating the error correction filtercoefficients for gain correction unit GC and phase correction unit PC.Rather, the responses shown in FIGS. 9a and 9b have to be converted tothe phase and gain correction responses shown in FIGS. 11a and 11bfrequency point by frequency point. The manner in which this isaccomplished is illustrated by the vector diagram shown in FIG. 10. InFIG. 10, the vectors correspond to signals developed at respectivelylabeled points of the direct conversion tuner shown in FIG. 5. Withreference to FIG. 10, the phase (d) and gain errors at each frequencypoint of the response shown in FIGS. 9a and 9b are used to calculate acoefficient for changing the magnitude of vector C and coefficient forchanging the magnitude of a quadrature vector F so that when theresultant vectors C' and F' are combined, vector G is formed which isequal in magnitude but opposite in phase to vector D, and thereforresults in cancellation when added to vector D. The desired phase andgain correction compensation responses shown in FIGS. 11a and 11bcorrespond to the multiplication factors for vectors C and F required toproduce vectors C' and F' for each sampled frequency on the frequencyaxis. The responses shown in FIGS. 11a and 11b are not symmetrical dueto interaction between gain and phase.

The coefficients for phase correction unit PC and gain correction unitGC can be computed from the responses shown if FIGS. 11a and 11b usingthe MatLab™ FIR2 program. It should be noted that the coefficients forfrequencies below second local oscillator frequency ω_(N) and the onesneeded for frequencies above second local oscillator frequency ω_(N)correspond respectively to the opposite sides of the responses shown inFIGS. 11a and 11b because the inverse positional relationship betweenthe desired components and the undesired components. This factor iseasily incorporated in the hardware by inverting the signal developed atpoint D as is indicated by inverter I as shown in FIG. 5, or bysubtracting the signal developed at point D from the signal developed atpoint G.

Once the filter coefficients for the selected channel have beencalculated they are stored for retrieval whenever the same channel isagain selected. As a result, the receiver is ready for reception when anew channel is selected without first calculating "new" filtercoefficients. Temperature affects the drift of components, such asinductors and capacitors, in the IF filter and therefor affects the gainand phase error responses. However, it has been found that it is notnecessary to perform the ripple gain and phase error measurements otherthan when the receiver is first turned on or perhaps when a new channelis initially selected. Accordingly, it is not necessary to interrupt thereception of a program. Measurement of the DC gain and phase errors isnot a problem because the received television signal can be continuouslymonitored without affecting the normal operation of the tuner.

It has been found that the location of the frequency, ω₀, of the firstlocal oscillator signals within the spectrum of the RF signal can beused to optimize the operation of the direct conversion tuner. Forexample, the frequency of the first local oscillator signals shoulddesirably between 1.7 and 2 MHz above the picture carrier (see FIG. 4a)for NTSC television signals, and between 2 and 2.8 MHz above the picturecarrier for PAL television signals. The frequency of the first localoscillator signals can be set in accordance with the channel number, forexample, by a phase locked loop tuning control system (sometimes calleda "frequency synthesizer"). In that case, microcomputer MC may be usedto control the frequency of the frequency determining programmabledivider of the phase locked loop. An automatic fine tuning (AFT)arrangement responsive to the frequency of the picture carrier may beused to maintain the frequency of the first local oscillator signals atthe desired frequency. The frequency of the picture carrier may bemeasured by microcomputer MC during the DC error measurement operationwhen the signals developed at points A and B are sampled.

The direct conversion tuner shown in FIG. 1 has been described so farwith respect to tuning of analog television signals in which picture,sound and color subcarrier signals are modulated on RF carriers inaccordance with a conventional television standard such as NTSC, PAL orSECAM. However the direct conversion tuner is also useful, and may infact be even more useful, for tuning digital television signals, such asHDTV (high definition television) signals. As earlier noted, HDTVsystems make full use of the available channel spectrum, have only asmall guard band of a few hundred kilohertz (KHz) between channels, andrequire a tuner response which is flat to the channel edges but which isvery steep at the edges for adequate adjacent channel rejection. Adirect conversion tuner is particularly well suited to such a HDTVenvironment because it has a low IF frequency range which allows the useof simple and effective filters. An IF filter with a sharp "cutoff" anda large "stop-band" region is much easier to obtain at low frequenciesthan at the conventional IF frequencies (38 MHz and higher).

While the direct conversion tuner described so far is well suited fortuning digital television signals, certain modifications need to beincorporated because discrete carriers which can be used to measure theDC gain and phase errors are usually not transmitted in a digitaltelevision system. However, it has been found that the spectra ofdigital television signals, which are typically flat and havesimilarities to random noise, may be used to accurately measure the DCerrors. The use of the spectra of the digital television signals tomeasure the DC errors may require more samples to be used than whenpicture or sound carriers are used. For example, samples taken over tento twenty television lines may be required. In a direct conversion tunerfor tuning digital television signals, the same method of measuring theripple gain and phase errors using a plural frequency reference signalpreviously described with respect to the direct conversion tuner fortuning analog television signals is used.

Even if the spectra of the received digital television signals are notflat for a particular digital television system, the spectra can stillbe used to measure the DC gain and phase difference errors, providedthat the shape of the spectra is to modify filter coefficients.

Digital television signals are more robust than analog televisionsignals and therefore the generation of unwanted spurious frequencycomponents is less critical. It is therefore not necessary to considerthe small asymmetric errors and larger tolerances for gain and phaseerrors can be accepted.

In a television receiver which is capable of processing both analogtelevision signals and digital television signals, a single directconversion tuner may be used to tune both the analog and digitaltelevision signals. The flow chart shown in FIG. 12 summarizes theoperation of the direct conversion tuner which has been previouslydescribed, and additionally indicates its operation in a dual modetelevision receiver. As is indicated in the flow chart, after thereceiver has been energized or a desired channel is selected, thepreviously stored gain and phase equalization data for the selectedchannel is retrieved from memory and coupled to gain correction unit GCand phase correction unit PC, the ripple error measurements are made.Thereafter, the detection of the presence or absence of a picturecarrier causes the selection of either an analog television signalbranch or a digital television signal branch, respectively, of theprogram for measuring the DC errors and calculating the filtercoefficients.

While the invention has been described in terms of as specificembodiment, it is contemplated that modifications will occur to thoseskilled in the art. For example while individual gain and phasecorrection units are utilized in the embodiment, a single digital filtermay be provided to provide both gain correction and phase correction.Such a filter may be constructed in either FIR (finite impulse response)or IIR (infinite impulse response) form. In addition, while gaincorrection unit GC and phase correction unit PC are included in the samechannel in the embodiment, one of the units may be included in onechannel and the other may be included in the other channel. Further,while two analog-to-digital converter are utilized in the embodiment, itis possible to use a single ADC which is multiplexed to sample thesignals developed at the measurement points of the two channels. Stillfurther, while the use of the picture and sound carriers has beendescribed with respect to the DC and asymmetry error measurements in adirect conversion tuner for tuning analog television signals, othercomponents may also be utilized. For example, the color subcarrier maybe used in place of sound carrier in the asymmetry error measurement.Even still further, while the reference signal is inserted after thefirst mixers in the embodiment, the reference signal can be inserted atother locations such as in the RF stage. In the same vain, while themeasurement points of the embodiment are located before the secondmixers because the second mixers are implemented in digital form,different measurement points, such as ones located after the secondmixers, may be used. Moreover, while a direct conversion tuner includingthe automatic gain and phase equalization provisions which have beendescribed is particularly well suited for tuning television signals, itis also useful for tuning other types of communications signals. Theseand other modifications are intended to be within the scope of thefollowing claims.

I claim:
 1. In a receiver, tuning apparatus for tuning a selected one ofa plurality of RF signals received at an RF input to produce an outputsignal at an output, comprising:first and second channels each having aninput and an output and, in the order named, a first mixer stage, afilter stage and a second mixer stage coupled between said input andoutput; said inputs of said first and second channels being coupled tosaid RF input; a summing unit having first and second inputs and anoutput, said outputs of said first and second channels being coupled torespective inputs of said summing unit, and said output of said summingunit being coupled to the output of said tuning apparatus; means forproviding first local oscillator signals of the same frequency but ofquadrature phases to respective ones of said first mixer stages; saidfrequency of said first local oscillator signals being located withinthe frequency spectrum of said selected RF signal; means for providingsecond local oscillator signals of the same frequency but of quadraturephases to respective ones of said second mixer stages; said frequency ofsaid second local oscillator signals being located above the passbandsof the respective said filter stages; means for monitoring first andsecond signals produced at respective points within said first andsecond channels; means for adjusting the relative gain and the phaseshift of said first and second channels prior to summing of said outputsof said first and second channels in said summing unit; said gain andphase shift adjusting means comprising respective individual units andbeing included in the same one of said first and second channels; meansfor automatically controlling said gain and phase shift adjusting meansto reduce the differences between the relative amplitudes and the phasesof said first and second channels in response to the relative amplitudesand phases of said first and second signals, wherein said gain adjustingmeans is coupled in cascade with an additional summing network betweensaid second mixer and said output of said one channel; and said phaseshift adjusting means comprises an additional mixer and an additionalgain adjusting means coupled in cascade between said filter stage andsaid output of said one channel; said additional mixer receiving asignal having the same frequency as the frequency of said second localoscillator signal of said one channel but of quadrature phase.
 2. In areceiver, tuning apparatus for tuning a selected one of a plurality ofRF signals received at an RF input to produce an output signal at anoutput, comprising:first and second channels each having an input and anoutput and, in the order named, a first mixer stage, a filter stage anda second mixer stage coupled between said input and output; said inputsof said first and second channels being coupled to said RF input; asumming unit having first and second inputs and an output, said outputsof said first and second channels being coupled to respective inputs ofsaid summing unit, and said output of said summing unit being coupled tothe output of said tuning apparatus; means for providing first localoscillator signals of the same frequency but of quadrature phases torespective ones of said first mixer stages; said frequency of said firstlocal oscillator signals being located within the frequency spectrum ofsaid selected RF signal; means for providing second local oscillatorsignals of the same frequency but of quadrature phases to respectiveones of said second mixer stages; said frequency of said second localoscillator signals being located above the passbands of the respectivesaid filter stages; means for monitoring first and second signalsproduced at respective points within said first and second channels;said means for monitoring said first and second signals including firstand second analog-to-digital converters respectively; said first andsecond digital-to-analog converters being coupled between respectiveones of said filters and said second mixer stages of said first andsecond channels; said second mixer stages of respective ones of saidfirst and second channels being digital mixers; means for adjusting therelative gain and the phase shift of said first and second channelsprior to summing of said outputs of said first and second channels insaid summing unit; means for automatically controlling said gain andphase shift adjusting means to reduce the differences between therelative amplitudes and the phases of said first and second channels inresponse to the relative amplitudes and phases of said first and secondsignals; said means for automatically controlling said gain and phaseshift adjusting means including a microcontroller operating underprogram control and being coupled to said analog-to-digital converter,wherein said automatic controlling means includes means for generating areference signal and inserting it into each of said first and secondchannels after respective ones of said first and second mixers. 3.Tuning apparatus for tuning to a selected one of a plurality of RFsignals received at an RF input to produce an output signal at anoutput, comprising:first and second channels each having an input and anoutput and, in the order named, a first mixer stage, a filter stage anda second mixer stage coupled between said input and output; said inputsof said first and second channels being coupled to said RF input; asumming unit having first and second inputs and an output, said outputsof said first and second channels being coupled to respective inputs ofsaid summing unit, and said output of said summing unit being coupled tothe output of said tuning apparatus; means for providing first localoscillator signals of the same frequency but of quadrature phases torespective ones of said first mixer stages; said frequency of said firstlocal oscillator signals being located within the frequency spectrum ofsaid selected RF signal; means for providing second local oscillatorsignals of the same frequency but of quadrature phases to respectiveones of said second mixer stages; said frequency of said second localoscillator signals being located above the passbands of the respectivesaid filter stages; means for monitoring first and second signalsproduced at respective points within said first and second channels;means for adjusting the relative gain and the phase shift of said firstand second channels prior to summing of said outputs of said first andsecond channels in said summing unit; and control means coupled to saidmonitoring and adjusting means for inserting a reference signal intoboth channels for automatically controlling said gain and phase shiftadjusting means to reduce the differences between the relativeamplitudes and the phases of said first and second channels in responseto the relative amplitudes and phases of said first and second signals.4. The tuning apparatus recited in claim 3, wherein:said referencesignal has a plurality of frequencies; and said control means isresponsive to said first and second signals as affected by saidreference signal at each of said pluralities of said frequencies.
 5. Thetuning apparatus recited in claim 3, wherein:said reference signal has aplurality of frequencies and said automatic controlling means isresponsive to said first and second signals as affected by saidreference signal at each one of said pluralities of said referencesignal.
 6. The tuning apparatus recited in claim 3, wherein:said gainadjusting means is coupled in cascade with an additional summing unitbetween said second mixer stage and said output of said one channel;said phase shift adjusting means comprises an additional mixer stage andan additional gain adjusting means coupled in cascade between saidfilter stage and said output of said one channel; and said additionalmixer stage receives a signal having the same frequency as the frequencyof said second local oscillator signal of said one channel but ofquadrature phase.